The present invention relates to a radar apparatus mounted on a vehicle such as an automobile, and used to constitute, for instance, a vehicle-to-vehicle safety distance warning system. More specifically, the present invention is directed to a peak value correction of an amplitude level, capable of correcting a peak value of an amplitude level of a detected spectrum as a correct value.
As this sort of radar apparatus, an FMCW radar apparatus is known. That is, since a transmitting/receiving common antenna is employed, a compact FMCW radar apparatus can be constructed and thus, can be easily mounted on an automobile. FIG. 4 is a block diagram for representing an arrangement of a conventional on-vehicle radar apparatus. In FIG. 4, reference numeral 1 indicates an oscillator, reference numeral 2 shows a power divider, reference numeral 3 represents a transmitter amplifier, and reference numeral 4 denotes a circulator. Also, reference numeral 5 indicates a transmitting/receiving common antenna, and this antenna is arranged by an electromagnetic radiator 51 and a reflection mirror 52. Furthermore, reference numeral 6 indicates a target object, reference numeral 7 indicates a receiver amplifier, reference numeral 8 represents an IQ detecting mixer, reference numeral 9 shows a filter, and reference numeral 10 indicates an AGC amplifier. Further, reference numeral 11 represents an A/D converter, reference numeral 12 shows a signal processing apparatus, reference numeral 13 indicates an antenna scanning motor, and reference numeral 14 represents an handle angle sensor.
Next, operations of the conventional radar apparatus with employment of the above-described arrangement will now be explained. The signal processing apparatus 12 outputs a linear voltage signal for an FM modulation. In response to this FM-modulating voltage signal, the oscillator 1 produces an FM-modulated electromagnetic wave. This electromagnetic wave is divided into two wave portions by the power divider 2. One divided electromagnetic wave portion is entered into the IQ detecting mixer 8. After the other divided electromagnetic wave portion is amplified by the transmitter amplifier 3, the amplified electromagnetic wave portions radiated via the circulator 4 from the transmitting/receiving common antenna 5 to the space. The electromagnetic wave which is radiated as a transmission electromagnetic wave from the transmitting/receiving common antenna 5 to the space is reflected from the target object 6, and then is entered into the transmitting/receiving common antenna 5 as a reception electromagnetic wave having a delay time "Td" with respect to the transmission when the target object 6 owns a relative speed, the reception electromagnetic wave having a Doppler shift "fd" with respect to the transmission electromagnetic wave is inputted to the transmitting/receiving common antenna 5. After the electromagnetic wave received by the transmitting/receiving common antenna 5 is amplified by the receiver amplifier 7, the amplified electromagnetic wave is mixed with the electromagnetic wave produced from the oscillator 1 by the IQ detecting mixer 8, so that a beat signal corresponding to both the delay time "Td" and the Doppler shift "fd" is outputted. The resulting beat signal is filtered by the filter 9, and the filtered signal is amplified by the AGC amplifier 10, and thereafter, the amplified signal is entered into the A/D converter 11. Based upon the A/D-converted beat signal, the signal processing apparatus 12 calculates a distance measured from the target object 6 and a relative speed.
Next, a description will now be made of a method for calculating a distance and a relative speed. FIG. 5 is an explanatory diagram for explaining an example of a method for calculating a distance and a relative speed by a conventional on-vehicle radar apparatus. In FIG. 5, a transmission electromagnetic wave is FM-modulated by a frequency sweeping bandwidth "B" and a modulation period "Tm". A reception electromagnetic wave owns delay time "Td" defined by such that the transmission electromagnetic wave is reflected from a target object 6 located at a distance "R" and then the reflected transmission electromagnetic wave is entered into the transmitting/receiving antenna 5. Also, when the target object 6 owns a relative speed "V", a reception electromagnetic wave is Doppler-shifted by "fd" with respect to a transmission electromagnetic wave. At this time, both a frequency difference "Fbu" between a transmission signal and a reception signal when a frequency is increased, and another frequency difference "Fbd" between a transmission signal and a reception signal when a frequency is decreased are outputted as a beat signal from an IQ detecting mixer 8. This beat signal is acquired via an A/D converter 11 into a signal processing apparatus as data. This acquired beat signal is processed by way of the FFT (Fast Fourier Transform) so as to obtain the frequency differences "Fbu" and "Fbd", and also a peak value "M" of amplitude levels thereof, as shown in FIG. 6. It should be understood that the peak value of "M" is a value equivalent to a reception strength, and will be referred to as a "reception strength" hereinafter.
A method for obtaining the frequency differences "Fbu" and "Fbd", and also the reception strength "M" will now be summarized as follows: That is, when the FFT process operation is carried out, the amplitude signals with respect to the respective abscissa time and ordinate time can be converted into the amplitudes of the frequency components with respect to the respective abscissa frequency and ordinate frequency. In the case that the frequency difference "Fbu" and the reception strength "M" are acquired, generally speaking, such a peak point where a level of amplitude becomes a peak is found out, and an amplitude level value of this peak and a frequency value thereof are assumed as the reception strength "M" and the frequency difference "Fbu". This frequency acquisition is similarly applied to another frequency "Fbd". In general, the reception strengths of the frequency differences "Fbu" and "Fbd" are identical to each other, and become "M".
Based upon the above-described items "Fbu", "Fbd", "Tm", and "B", the light velocity "C(=3.0.times.10.sup.8 m/s)", and a wavelength ".lambda." of a carrier wave (if a basic frequency of a carrier wave is defined as f.sub.0 =77 GHz, then a wavelength ".lambda." is given as .lambda.=4.0.times.10.sup.-3 m), the distance "R" and the relative speed "V" of the target object 6 are calculated by the below-mentioned formulae (1) and (2): EQU R=(TmC/4B).times.(Fbu+Fbd) (1) EQU V=(.lambda./4).times.(Fbu-Fbd) (2)
Also, in the case that a plurality of target objects are located, based upon a plurality of frequency differences "Fbu" between transmission signals and reception signals when a frequency is increased, and a plurality of frequency differences "Fbd" between transmission signals and reception signals when a frequency is decreased, both "Fbu" and "Fbd" of the same object are selected. Then, the distance "R" and the relative speed "V" are obtained from the above-described formulae (1) and (2).
Next, operations of the IQ detecting mixer 8 will now be explained in detail. In FIG. 4, the electromagnetic wave produced from the oscillator 1 is distributed to the power divider 2, and is further subdivided by 1/2 into two electromagnetic wave portions by the power divider P/D at the input unit of the IQ detecting mixer 8, and then, these two electromagnetic wave portions are entered as LO (local) signals into mixers 81 and 82. Also, the received electromagnetic wave is amplified by the reception amplifier 7, and thereafter, the amplified electromagnetic wave is subdivided by 1/2 into two electromagnetic wave portions by the power divider P/D. One subdivided electromagnetic wave portion is directly entered into the mixer 81. The other subdivided electromagnetic wave portion is entered via a 90-degree signal line (1/4 wavelength) to the mixer 82. In this case, a beat signal outputted from the two mixers 81 and 82 outputs an In-phase component "I" and also a Quadrature component "Q" having a phase difference of 90 degrees with respect to the In-phase component "I". The IQ components derived from these mixers 81 and 82 are sampled by the A/D converter 11. Then, the I component and the Q component are processed as a real number portion and an imaginary number portion by the complex FFT processing operation.
When the complex FFT processing operation is carried out, it is possible to judge as to whether or not the frequency component of the spectrum is equal to a positive component based upon the positive/negative value of the phase difference by 90 degrees. As indicated in FIG. 6, only one spectrum appears after the IQ components are FFT-processed. In other words, when the FFT processing operation is carried out by employing the normal mixer, one pair of spectrums whose positive/negative frequency components are inverted appear on the frequency axis, so that it is not possible to judge as to whether the frequency component is equal to a positive value or a negative value. To the contrary, when the FFT processing operation is carried out by employing the IQ detecting mixer 8, as represented in FIG. 6, since only one spectrum appears after the IQ components are FFT-processed, it is possible to judge as to whether the frequency component of the FFT-processed IQ components is equal to a positive value or a negative value. As a consequence, it is possible to decrease an artifact, or a false image which is produced by mistakenly combining the frequency difference "Fbu" with the frequency difference "Fbd" in such a case that a plurality of targets are detected by employing the IQ detecting mixer rather than using the normal mixer.
Next, a description will now be made of a method for calculating a direction of the target object 6 by the signal processing apparatus 12 from the reception strength "M". As the conventional method for calculating the direction of the target object, the following typical methods have been disclosed, i.e., the mono-pulse method, the sequential lobbing method, and the conical scanning method, for example, in the Examined Japanese Patent Application Publication No. Hei 7-20016. In this case, the sequential lobbing method is described. This sequential lobing method is equivalent to the method disclosed in Japanese Laid-open Patent Application No. Hei-7-92258, namely, such an angle measuring method capable of measuring an angle over a wide range while using a normalized difference between reception strengths of two radar beams having different axes.
The sequential lobbing method is summarized as follows. That is, after a distance, a relative speed, and a reception strength M have been measured along a preselected direction ".theta.1", the signal processing apparatus 12 actuates the motor 13 so as to move the transmitting/receiving common antenna 5 along a next direction ".theta.2", and then, similarly, measures a distance, a relative speed, and a reception strength "M2". The signal processing apparatus 12 selects the same distance data and the same relative speed data from the data detected along these plural directions, and can basically measure an angle by checking as to whether or not the reception strength M1 is higher than the reception strength M2.
It should be understood that symbol ".theta." indicates an angle, in which a front direction of a vehicle is set to "0.degree.", a forward right oblique direction of the vehicle is set to a "positive angle", and a forward left oblique direction of the vehicle is set to a "negative angle".
Concretely speaking, both a summation pattern S(.theta.) and a difference pattern D(.theta.) are calculated from an antenna beam pattern B1(.theta.) and another antenna beam pattern B2(.theta.) along two predetermined directions ".theta.1" and ".theta.2" by the below-mentioned formulae: EQU S(.theta.)=B1(.theta.)+B2(.theta.) (3) EQU D(.theta.)=B1(.theta.)-B2(.theta.) (4)
Next, DS(.theta.) of the following formulae which is normalized by S(.theta.) is obtained= EQU DS(.theta.)=D(.theta.)/S(.theta.) (5)
It should be noted that DS(.theta.) is a monotone increase, or a monotone decrease with respect to ".theta." within a half-value width ".theta.s" of S(.theta.).
Next, while a center between .theta.1 and .theta.2 along two predetermined directions is set as ".theta.o" and a half-value width of S(.theta.) is set as ".theta.s", both an angle ".theta.n" normalized by ".theta.s" and an inclination "K" of DS(.theta.) in the vicinity of .theta.n=0 are calculated by the following formulae= EQU .theta.n=(.theta.-.theta.o)/.theta.s (6) EQU k=DS(.theta.)/.theta.n (7)
Also, DS acquired from a monitoring result is calculated based on both the reception strength M1 and the reception strength M2 by the following formulae: EQU DS=(M1-M2)/(M1+M2) (8).
As a result, based upon the precalculated .theta.s, K, .theta.o, and also DS acquired from the monitoring operation, the angle ".theta." may be calculated by the below-mentioned formula (9):
.theta.=(.theta.s/k).multidot.DS+.theta.o (9).
A relative position of a car driven ahead can be grasped based on the above-measured distance and also angle up to the target object. Also, when a curvature of a road can be grasped from the handle angle sensor 14, a position of a lane along which the own car is driven (lane width is predetermined as 3.5 m). As a result, it can be seen whether or not the car driven ahead is traveled on the same lane as the own car. A judgement is made in this manner as to whether or not the target object corresponds to the car driven ahead which is traveled on the same lane as the own car. Based upon the judgement result, the signal processing apparatus issues the vehicle-to-vehicle safety distance warning notification and also executes the forward-vehicle-following drive so as to keep the safety vehicle-to-vehicle distance.
Since two sets of output signals from the IQ detecting mixer 8 are entered via the A/D converter 11 into the signal processing apparatus 12 so as to be FFT-processed in the above-explained conventional radar apparatus, when the unbalance IQ amplitude value occurs and the IQ phase error occurs in the I-channel signal and the Q-channel signal, which are inputted into the A/D converter 11, as represented in FIG. 7, false spectrums will appear in the frequencies of the spectrums, the symbols of which are inverted. Also, the larger the degrees of the IQ phase error and the unbalance IQ amplitude values are increased, the larger the amplitude level of the false spectrum becomes. The causes of the phase error/unbalance amplitude value may be conceivable from the correctness of the 90-degree signal path of the IQ detecting mixer 8, the balance of the power divider P/D of the IQ detecting mixer 8, the fluctuations in the conversion losses of the two mixers 81/82 employed in the IQ detecting mixer 8, the temperature characteristic of the IQ detecting mixer 8, and the multiplexing reflection. In principle, although the phase error and the unbalance amplitude value of the I, Q-channel signals caused by the above-described items may be reduced, these unbalance reasons cannot be completely solved in view of cost, and further under drive conditions of automobiles. Under such a reason, there are problems that the amplitude of the true spectrum is reduced, the detection performance of the true spectrum is deteriorated, and the angle-measuring calculation error happens to occur due to the amplitude error of the true spectrum.